Measuring circuit and measuring method for a capacitive touch-sensitive panel

ABSTRACT

A measuring circuit connectable to a capacitive touch-sensitive panel, the panel including a plurality of sense electrodes and optionally a common guard electrode, adapted to measure variations in the instantaneous electric capacity of the sense electrodes in response to proximity to conductive bodies, wherein the sense electrodes are biased at a fixed voltage relative to the common guard electrode, the measuring circuit comprising: a power management integrated circuit comprising a voltage source generating a modulation voltage that is available at a guard terminal of the power management integrated circuit that is in electric connection with the guard electrode; one or more slave integrated circuits, each connected to a plurality of sense electrodes and comprising a Capacity-to-Digital converter or a plurality of Capacity-to-Digital converters that are operatively arranged for generating digital measure codes representing the instantaneous electric capacity of sense electrodes; a means for varying the frequency of the modulation voltage.

REFERENCE DATA

The present application claims priority from U.S. provisional patentapplication 62/026,178 of Jul. 18, 2014, in the name of the SemtechCorporation, Camarillo, Calif., the contents whereof are herebyincorporated by reference in their entirety.

FIELD OF THE INVENTION

The present invention relates to a measuring circuit and measuringmethod for a capacitive touch-sensitive panel; and in particularly to ameasuring circuit and measuring method for a capacitive touch-sensitivepanel involving varying the modulation frequency of a modulating voltageso as to smooth the peaks in the transfer function.

DESCRIPTION OF RELATED ART

FIG. 1 describes a known technique for measuring a grounded capacitorCin, which might be part of a touch-sensitive panel, or of a proximitydetector: it consists into varying the voltage of the capacitiveelectrode and detecting the corresponding charge variation across Cin.This is generally achieved by tying the capacitive electrode to thenegative input (virtual ground) of a charge amplifier with a capacitorCfb in feedback. The voltage variation on the input capacitor isachieved by applying a well-defined voltage variation on the positiveinput of the amplifier, as the negative input will track the positiveone by feedback. Since the current across the capacitor Cin only flowstowards Cfb (the amplifier having high impedance inputs), the chargevariation across Cin (and thus the value itself of Cin) may bedetermined from the voltage variation across feedback capacitor Cfb.This voltage variation can be measured directly in the analog domain,processed, or converted into the digital domain.

One drawback of this technique is its extreme sensitivity to anyparasitic capacitor Cpar between electrode input node and ground, and inparticular to the parasitic capacitors related to input pads,protections and parasitic capacitors of input amplifier, parasiticcapacitors to supply voltages. Indeed, these parasitic capacitors maynot be distinguished from the capacitor to be measured and thus affectthe measurement result.

Patent FR 2 756 048 describes techniques for measurement of a groundedcapacitor, as typically used for proximity detection. The advantage ofthese techniques lies in their precision and in that they are quiteinsensitive to parasitic capacitors. This is achieved by varying withrespect to ground not only the voltage of the capacitive electrode butall the voltages of the measuring circuitry. All the voltages vary inthe same way as the voltage of the capacitive electrode such that thevoltage across the parasitic capacitors does not change. To this end,all the input circuit or charge amplifier is referred to a localreference potential, also named a local ground (typically the substrateof the measurement circuit), which is caused to vary with respect to theglobal ground by some excitation circuit, such the voltage source thatgenerates the varying voltage Vin, see FIG. 2. The local ground(floating voltage VF) is thus floated with respect to the global(external) ground. The readout circuit is supplied by floating positiveand negative supplies that are referenced to local ground. Frommeasurement circuit point of view, “only” the external ground voltage ischanging, all the internal circuitry being referred to floating voltage.Hence the measurement is insensitive to parasitic internal capacitors.

The capacitor Cin to be measured may be far from the measurementcircuitry, however, so any parasitic capacitor between the wireconnecting Cin to measurement circuit would be added to the measuredcapacitor. To avoid this error, the wire connecting Cin to themeasurement circuitry may be uncoupled from the external ground by usinga guard electrode. This guard electrode must then be connected to theinternal or floating ground VF or to a node biased at a constant voltagewith respect to VF, such that the capacitor between capacitive electrodeand guard remain biased at a constant voltage and does not affect themeasurement result. For this reason, the measurement circuitry has aguard output tied to internal ground VF or biased at a constant voltagewith respect to it, and the guard of the wire between capacitor andmeasurement circuit should be tied to this output of the measurementcircuit, see FIG. 3.

BRIEF SUMMARY OF THE INVENTION

The essential features of the present invention are recited in theindependent claims of this application. Further optional, favorable,features of other embodiments are mentioned in the dependent claims.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be better understood with the aid of the descriptionof an embodiment given by way of example and illustrated by the figures,in which:

FIGS. 1 to 3 show, schematically, known circuits used in capacitymeasurements;

FIG. 4 illustrates a LCD panel overlaid by a transparent guardelectrode, above which are placed a plurality of conductive transparentpixels, and a part of a capacity measuring apparatus;

FIG. 5 shows the building blocks of a possible embodiment of the presentinvention;

FIGS. 6 and 7 illustrate in a schematic fashion two possible manners ofaveraging the variation of the output signal of the charge amplifierover different modulation cycles;

FIG. 8 illustrates a third circuit for averaging the variation of theoutput signal of the charge amplifier that makes use of a delta-sigmamodulator;

FIG. 9 illustrates a circuit wherein a perturbing voltage is perturbingthe voltage on an electrode of the capacitor Cin to be detected;

FIG. 10 illustrates a modulation voltage, output voltage and samples,which are obtained in the circuit shown in FIG. 9 when the relevantsignals are sampled at discrete times;

FIG. 11 show curves which depict the transfer function for perturbingsignals when the modulation voltage has a fixed modulation frequency,and when the modulation voltage has a frequency varied between fmod1,fmod2, fmod3 and fmod4, and curve showing an average;

FIGS. 12a to 12e shows some examples of how the modulation frequency ofthe modulation voltage can vary.

DETAILED DESCRIPTION OF POSSIBLE EMBODIMENTS OF THE INVENTION

In a display application, typically for smartphones or tablets, thecapacitive electrodes are placed on top of an LCD display and thecapacitances to be measured are between these top electrodes andexternal ground, through the finger approaching the screen.

Only the capacitance on the upper side with respect to fingers is ofinterest, however, while the capacitance with respect to LCD andparasitic signals from LCD is not useful to detect finger's proximityand, indeed, the activity of the LCD is liable to inject unwantedcharges in the readout circuit through the parasitic capacitors, whichcould false the output of the proximity detector. For this reason, aconducting guard layer is inserted between the capacitive electrodes andthe LCD display. This conducting guard layer should also be tied to theguard output of the measurement circuitry, or held at a constant voltagerelative to the guard output, as it was the case for the guard of thewires between touch screen and measurement circuitry.

Such an arrangement is exemplified in FIG. 4 in which a LCD panel 200 isoverlaid by a transparent guard electrode 30, above which are placed aplurality of conductive transparent pixels 25 that are connected to aplurality of Capacity-to Digital Converter included in a readout circuit120. Each CDC includes a charge amplifier 126. Since the guard electrodecan be regarded as an equipotential surface, it provides an effectiveelectrostatic screen, and unwanted interferences that may come from theLCD 200 are effectively screened by the guard potential and do not reachthe CDCs.

As discussed above, the readout circuit includes a variable voltagesource 80 that generates a floating reference potential 85 that isconnected to the guard potential 30 and to the non-inverting inputs ofthe charge amplifiers 126 of the CDC. The variable voltage source couldgenerate a square wave signal, as it is shown in the figure, or also acontinuous variable waveform, like for example a sinusoid, withoutleaving the spirit and scope of the present invention. A squareexcitation may be preferable in discrete-time systems, while continuousexcitation could be favored by implementation that make heavy use ofanalog processing.

In this configuration, the CDC stages have low-impedance virtual groundinputs and the pixel electrodes 25 are essentially held at the potential85 of the floating reference potential. The amplitude of the signal atthe outputs Vout_1, Vout_2 . . . Vout_N is proportional to therespective capacities towards ground Cin_1, Cin_2, . . . , Cin_N, seenby the electrodes 25. Importantly, the voltage across parasiticcapacitors 212, which are connected between the guard electrode 30 andthe pixels 25 is constant, hence these parasitic elements do notcontribute to the readout.

The guard electrode 30 is useful, as explained above, for reducing thepick-up of unwanted signal generated by the LCD screen by the pixelelectrodes, but it is not an essential feature of the invention, whichcould function even if the guard electrode 30 were omitted, provided thenon-inverting inputs of the integrators 126 are connected to thefloating reference potential 85. This simplified configuration wouldlead to a thinner touch-sensitive panel and may indeed be advantageousin specific circumstances.

It is noted also that, despite the guard electrode 30, thetouch-sensitive electrodes 25 will pick up a fair deal of unwantedsignals that have nothing to do with the finger capacity, either fromthe LCD 200 or from other interfering sources: firstly, the guardelectrode 30 itself has a finite conductivity and cannot be strictlyequipotential; secondly because the guard 30 is limited in size, andfinally because the upper side of the electrodes 25 is not shielded atall, nor can it be, and is liable to receive interfering signals fromthe mobile phone itself, or from any source in proximity.

The drawing show that the reference potential 85 for the chargeamplifier 126 is determined by the modulated voltage source 80 in thatnon-inverting inputs are directly tied to one of its terminals. Thepresent invention is not limited to this structure, however. Thereference potential 85 could be determined indirectly by the voltagesource by a buffer amplifier, by a non-represented voltage sourceservoed or synchronised to the voltage source 80, or by any otherappropriate means.

The circuit for measuring the external grounded capacitor includesseveral building blocks, represented in FIG. 5:

-   -   The excitation voltage source 80, used to generate the floating        node VF, or local ground, varying with respect to the global or        external ground.    -   The acquisition and measurement circuitry for measuring the        charge variation across the capacitor to be measured, and        producing a signal or, preferably, a digital code that        represents this capacity. Dependent on the number of capacitive        input pixels this circuitry may include a plurality of        independent capacity-to-digital converters 130, each referred to        floating ground VF. The converters 130 may comprise a charge        amplifier 126 (amplifier with feedback capacitor tied between        output and negative input, and with positive input tied to        floating ground VF (or guard)) and eventually other circuitry        for post processing, such as analog to digital converters 128,        filters, amplifiers, attenuators, or input multiplexers 127.    -   Generation of the supply voltages (V+, V−): since the converters        130 are referred to the floating ground, their active elements        should preferably be supplied with voltage sources that are        referenced to the floating ground rather than to the external        ground. The floating supply unit 175 produces the required        alimentations, from an external voltage supply vdd referred to        external ground. The floating supply 175 may include inductive        transformers, DC/DC converters of the boost or buck variety,        switched-capacitor circuits, or any other voltage conversion        scheme.    -   Generation of control and clock signals 182: many functions of        the acquisition circuits need to be synchronised with the        modulation signal applied between external ground and internal        or floating ground. In particular the detection of the charge        must be perfectly synchronous with the modulation signal.        Moreover, data coming from the acquisition units 130 need to be        transmitted outside the floating voltage domain.

In many applications, and particularly where touch screen and proximitydetections are concerned, a large number of capacitors must be measuredsimultaneously or successively. In order to track the movement of asliding finger, for example, the touch screen must be able to acquirethe values of all the capacitors 25 in a short time frame. Themeasurement circuitry may then include several acquisition chains oracquisition circuitry in parallel 130 for measuring a large number ofcapacitors. A multiplexer 127 may be added in front of each measurementcircuitry in order to address successively different input electrodes,one after the other, as illustrated in FIG. 5. The multiplexers 127 infront of the acquisition chains allow addressing several inputs insuccession by each acquisition chain, thereby reducing the number ofacquisition chains to implement on a chip.

The first block of the acquisition chain after the optional inputmultiplexer 127 is a charge amplifier 126. The output voltage of thecharge amplifier 126 exhibits a voltage variation which is synchronousand proportional to the voltage variation applied to the floating groundnode 85 (floating voltage VF or guard voltage). The variation in theoutput voltage of the charge amplifier 126 is also proportional to theinput capacitance (Cin_1, Cin_2, . . . , Cin_N) to be detected, and isthus the signal of interest. The purpose of the A/D 128 is thenprecisely to measure the output voltage variation of the chargeamplifier 126. This variation of the output voltage (Vout_1, Vout_2 . .. Vout_N in FIG. 4) of the charge amplifier 126 should preferably bemeasured in the floating supply domain, thus with respect to thefloating ground (guard, V+, or V−). It must be noted that modern touchuser interface require a very high resolution to the analog/digitalconverter 128. Converter with a resolution of 16 bits, or even higher,are not uncommon.

Since the signal generated by the charge amplifiers 126 is variable, itis preferably converted by a suitable detection means into a valueproportional to or indicative of its amplitude. The detection will beadapted according to the circumstances and especially dependent from thenature of the modulation signal. Preferably, however, the detection willbe synchronous with the excitation source, accepting selectively thecapacity signals that are synchronous with the excitation voltage, andrejecting unwanted disturbances that are not.

If for example the excitation source 80 is arranged to generate asinusoidal signal on the floating ground node VF, then the peak to peakamplitude of the sinusoidal signal at the outputs of the chargeamplifiers 126 can be detected by suitable demodulation schemes, forinstance by multiplying the output signal by the sinusoidal input signaland low pass filtering in order to eliminate the harmonics.

If on the other hand the excitation source 80 is arranged to generate asquare wave signal on the floating ground node VF, then the amplitude ofthe square wave output signal (Vout_1, Vout_2 . . . Vout_N) of thecharge amplifier 126 can be detected and measured, for example byquantifying the amplitude of the rising and falling edges. The risingand falling edges can be quantized separately, or summed up in theanalog domain and quantized. Different options are possible to carry outthe detection, both in the analog and in the digital domain.

However, regardless of how the input capacitor (Cin_1, Cin_2, . . . ,Cin_N) is converted into a voltage variation of output voltage (Vout_1,Vout_2 . . . Vout_N) of the charge amplifier 126, this measurement willbe affected by several noise i.e. perturbations, in particular but notexclusively:

-   -   Thermal noise of the circuit, basically due to resistors (4 k T        R noise), MOS transistors of amplifiers (4 k T/gm noise),        switches (leading to k T/C noise). This is wide band noise with        approximately flat noise spectral density (white noise), and    -   Interferers, parasitic signals coupling to the electrodes, for        instance due to 50/60 Hz power network, parasitic signals due to        battery chargers.

An effective way to attenuate these perturbations is to repeat themeasurement several times and at a rate considerably faster than thetarget frame rate, and to average this oversampled results in order tofilter out the perturbing signals and obtain a signal with less noise,and a lower bandwidth. The averaging can be a direct (un-weighted)averaging (sum of the samples divided by number of samples), or aweighted averaging (different samples having different weights whenaveraging).

In any case, this averaging corresponds to a low-pass filtering. Inorder to reduce the filter bandwidth and thus to eliminate most ofperturbations, it is desirable to average the measurement over a largenumber of cycles of the modulation signal (signal modulating thefloating ground). The bandwidth is then indeed inversely proportional tothe number of averaged modulation cycles.

Since, by averaging, the overall duration of the measurement isproportional to the number of modulation cycles times the period of themodulation cycle, and the bandwidth is inversely proportional to theoverall measurement time, there is a trade-off between the conversionrate or frame rate on one side, and the rejection of externalperturbation outside a narrow band on the other side. The cornerfrequence of the filter used cannot be lower than the frame rate.

The averaging of the variation of the output signal (Vout_1, Vout_2 . .. Vout_N) of the charge amplifier 126 over the different modulationcycles can be done in different ways. The filter used may be, within theframe of the invention, any suitable low-pass filter, be it an analogfilter, a discrete-time analog (switched-capacitor) filter, a digitalfilter, or a combination thereof.

A first exemplary solution is illustrated in FIG. 6; this first solutionincludes performing the averaging 160 of the output of the chargeamplifier 126 in the analog domain and then performing analog-digitalconversion 161. The drawback of this solution is that realizing thisaveraging on a narrow bandwidth with an analog low pass filter requireslarge capacitors and resistors, and thus a large area. Another drawbackis that it requires a high resolution ADC, also leading to a large area.

A second exemplary solution is illustrated in FIG. 7; this secondsolution comprises first performing analog to digital conversion 170 ofthe output of the charge amplifier 126, and then performing theaveraging in the digital domain within the digital averaging block 171.The advantage of this second solution is that that the digital filteringcan be realized efficiently, with low silicon area. However,disadvantageously a high resolution ADC is still required.

The third exemplary solution is to use a delta-sigma analog to digitalconverter (ADC converter) or an incremental ADC in order to perform theanalog to digital conversion. These types of ADC converters areespecially suited to this application thanks to their ability to achievea high resolution with a relatively low silicon area. These types ofconverters can as well include a digital postfilter that simultaneouslyperform the required averaging. An example of the third solution isillustrated in FIG. 8, using a first order sigma delta modulator 181;however it will be understood that higher order sigma delta modulatorsare possible. An incremental ADC would be similarly structured, however,in contrast to sigma-delta ADC that convert a waveform continuously, theincremental ADC converts a predetermined number of individual samplesand is then reset.

The variation in the output of the charge amplifier 126 at eachmodulation cycle, such as the peak to peak amplitude or the voltageedges, is first extracted by a detection unit 182. The output of thisdetection unit 182 is then integrated by an integrator 183 (which ispreferably a switched capacitor integrator operating also at themodulation frequency, although it will be understood that other types ofintegrators could be used). The output of this integrator 183 is thenconverted by a coarse quantifier 184 (comparator or bank of comparators)into small digital codes (1 bit or a very limited number of bits)produced at the same rate as the modulation rate. These codes are thenconverted back into analog by a digital-to-analog converter 185 andsubtracted from the input signal corresponding to the charge amplifier'soutput voltage variations (output of detection unit 182).

Due to the feedback loop to the input of the integrator 183, the outputcode is forced to match the input signal, at least for low frequencies.This means that at low frequencies the output code of the sigma deltaloop is a good representation of the charge amplifier's output voltagevariation. The output code is then filtered using a filter 186. Thus byfiltering the output code using a filter 186, or by averaging the outputcode from the sigma delta loop (averaging being in fact a particularcase of filtering), one obtains a digital output code which isrepresentative of the averaged (or low pass filtered) value of thecharge amplifier output voltage variation, and thus of the inputcapacitor. The averaging and ADC conversion are thus performedsimultaneously.

An advantage of this approach is that it does not require very largecapacitors in order to accumulate the signal corresponding to the chargeamplifier's output voltage variation. Indeed, as soon as the accumulatedsignal exceeds a given level, a quantity corresponding to the outputcode is subtracted by the feedback path. By this fact, a limited amountof signal is accumulated even after a large number of samples, as thefeedback loop manages to avoid saturation of the integrator. Thus, thisaccumulation does not require huge capacitors and silicon area.

Another advantage is that it is able to achieve a very high resolutionwith a very coarse quantifier, at the extreme with a simple comparatorproducing one bit at a time. Indeed, for instance by cumulating theoutput bit on 65536 cycles, it is possible to obtain a 16-bit resolutionoutput code. No high precision is required for the quantifier, as theerrors are compensated by the feedback loop.

In any case, whatever be the selected method for averaging, either inanalog or in digital or a mixed with a sigma delta modulator, the effectis to reduce the bandwidth for noise and interferers, and this bandwidthdecreases inversely proportionally with the number of averagedmodulation cycles.

The averaging method is very efficient rejecting the thermal noise byreducing its effective bandwidth. The improvement with respect tointerferers is not so straightforward, however. While most interferersare indeed strongly attenuated, giving a good overall improvement, someinterferers falling into the effective bandwidth may be attenuated veryweakly attenuated, or hardly at all. This is particular true forperturbations at frequencies equal to frequency of the modulationfrequency (fmod) or very close to it, or else eventually for frequenciesclose to the harmonics of the modulation frequency, especially theodd-order ones.

FIG. 9 illustrates a case wherein a perturbing voltage Vperturb isperturbing the voltage on the left electrode 190 of the capacitor Cin tobe detected. The voltage of this left electrode 190 should ideally bethat of the external ground; the perturbing voltage, however, shifts itsvoltage level to Vperturb. As the internal ground or floating ground ismoved by voltage source Vin (which is the modulating voltage signalwhich has a modulation frequency (fmod)) with respect to externalground, one can consider that the effective voltage applied on Cin withrespect to floating ground is thus Vin-Vperturb rather than the‘nominal’ Vin value. If the perturbing voltage has components atfrequencies very close to that one of the modulating signal Vin, thesecomponents may not be distinguished from the modulating signal duringthe measurement time, that means during the time corresponding to theaveraging of the different modulation cycles. In such case, thesecomponents may significantly corrupt the measurement result.

As an example of how the components of the perturbing voltage Vperturb,which have a frequency equal or close to the modulation frequency cancorrupt the measurement result, let us first consider a continuous timeapproach in which the modulation signal Vin is a pure sinusoidal signalof amplitude Vmod at modulation frequency fmod. This will result into anoutput voltage Vout of amplitude (Cin/Cfb)×Vmod at the output of thecharge amplifier 126 relative to floating ground 80. The value of thecapacitor Cin can then be extracted by demodulating this output signalof the charge amplifier 126, which means re-multiplying by a sinusoidsynchronous with Vin (same frequency fmod and same phase) and thenaveraging this demodulated signal in order to limit its bandwidth closeto DC. A perturbing signal Vperturb may then provide a significantcomponent close to DC at output even after averaging if its frequency isvery close to that of the demodulating signal such that its phase doesnot shift significantly (less than one period let us say) with respectto the demodulating signal during the measurement time.

We denote with Tobs the observation time or, equivalently, the timeduring which the output signal is averaged after demodulation, whichcorresponds to N×Tmod, with Tmod being the modulation period and N thenumber of averaged cycles. If the frequency difference between thedemodulating or modulating signal and the perturbing signal is less1/Tobs (|f-fmod|<1/Tobs), the perturbing signal cannot be efficientlydistinguished from the modulating signal and will thus significantlycorrupt the result. Frequencies further from the frequency fmod are lessproblematic, as they look more orthogonal to the modulation frequencyover the observation time.

Now, with respect to the circuit shown in FIG. 9, let us consider adiscrete time approach, such as typically a switched capacitor approach,in which the modulation signal Vin is a square wave signal. Forsimplicity's sake we will assume that the duty cycle of the modulationsignal is 50%, and that the output voltage Vout, which is also square,is sampled twice in each period of Vin, sufficiently far from the edgesin order to let the charge amplifier settle correctly, as is depicted inFIG. 10.

As it can be seen in FIG. 10 all the odd samples correspond for instanceto the low state and the even samples to the high state of themodulating signal Vin (or the opposite). The difference between odd andeven samples thus systematically corresponds to the signal(Cin/cfb)×Vmod and is thus representative of the capacitor to bedetected. The measurement then comprises extracting the average value ofthe difference between odd and even samples.

Computing the average difference between odd and even samplescorresponds to multiply all the samples alternately by +1 or −1, thus bya signal at fs/2, which corresponds to a demodulation in discrete time,with fs=2×fmod being the sampling rate. The sensitivity is maximum atfs/2=fmod, as the signal is then inverted between each sample, whichgives the maximum signal, as one computes the difference between odd andeven samples.

As in the continuous time approach, the measurement will then be verysensitive to perturbing signals whose frequency is very close to themodulation frequency fmod, as odd and even samples will then beperturbed differently, provided the frequency difference betweenperturbing signal and modulating signal is roughly below 1/Tobs.

In the same way, the measurement is not sensitive to DC or quasi DCsignals over the measurement time (f<1/Tobs), as the odd and evensamples are affected the same way so that they cancel each other whendemodulating by calculating the difference between odd and even samples.

Since the output signal Vout is sampled at frequency fs=2×fmod, thetransfer function for the perturbing signals is periodic with periodfs=2×fmod in the frequency domain. Therefore, as the signal isinsensitive to DC signals, because the signal is capacitively coupled,it will also be insensitive to all perturbing signals at k×fs=2k×fmod,that is to say, to signal having frequencies close to the even harmonicsof the modulating signals. Indeed, the sampling of such a signal may notbe distinguished from sampling of a DC signal.

For the same reason, as the signal presents a maximum sensitivity atfmod, it will also present maximum sensitivity for all odd harmonics offmod, thus frequencies f=fmod+k fs=(2k+1)×fmod. For all thesefrequencies, the transfer function will be essentially the same as forfmod, barring a slight attenuation due to parasitic or intentional RCfiltering. Thus in a discrete time (type switched capacitor approach),the rejection of perturbations will be rather poor around the modulationfrequency, and also for all the frequencies very close to its oddharmonics, at least as long as these harmonics are not attenuated byother filtering effects.

Other sampling schemes with a sampled or switched capacitor approach arepossible. For instance the rising and falling edges of the Vout signal,by sampling two values for each edge, one before the (rising or falling)edge and one after the edge, leading to 4 samples per modulation period.The advantage being that the noise involved by the reset of the chargeamplifier between two edges can be eliminated. Moreover, due to thehigher number of samples, theoretically more filtering can be performedin the discrete time. In practice, however, as more samples are takenper modulation period, the charge amplifier will need to be faster, witha larger bandwidth, and the consequence of this is that the measurementwill be sensitive to further harmonics, negating at least partially theadvantage of having more samples per period.

The approaches discussed with respect to FIGS. 9 and 10 presentgenerally good rejection of interferers due to averaging. They behavepoorly, however, when the interferers have frequencies very close to themodulation frequency fmod; in particular, the discrete time approach hasalmost no rejection for other frequencies, typically the odd harmonicsof the modulation frequency fmod, which are aliased into the modulationsignal baseband. The present invention aims to obviate or mitigate thisproblem.

In an embodiment of the present invention the modulation frequency fmodis varied when averaging the data over several modulation periods, inorder to smooth the peaks in the transfer function. This effect ofsmoothing the peaks of the transfer function by varying the modulationfrequency is depicted in FIG. 11.

In FIG. 11 the top-most curve shows what would be the transfer functionfor perturbing signals when the modulation voltage has a fixedmodulation frequency fmod; it can be seen that peaks 211 occurs aroundmodulation frequency fmod and typically at all the odd order harmonicsof the modulation frequency 3×fmod, 5×fmod, 7×fmod.

The four curves below the top-most curve show how this transfer functionwould change when varying the modulation frequency fmod slightly aroundits nominal value to get a frequency fmod_i (i=0 . . . 4 in thisexample); All these four curves are obtained by compressing or expandingthe first curve along the frequency axis by a factor fmodi/fmod.

Now, let us assume that during the overall acquisition of thecapacitors, the modulation frequency fmod is varied between fmod1,fmod2, fmod3 and fmod4, each of these frequencies being used for 25% ofthe modulation cycles. As a first approximation, one could expect thatthe resulting transfer function would approximately look like theaverage of the four previous curves, as shown in the lowest curve (theaverage curve). As can be seen the peak of the amplitude of the peaks214 in the average curve are strongly attenuated (in this case roughlyby a factor of 4), but the bandwidth is extended so that the bandwidthavailable for the perturbing signal is extended. Consequently, althoughthe measurement becomes more sensitive to perturbing signal lying in awider band, the averaged response function does not exhibits aspronounced peaks for any given frequency as the individual ones, and itsworst-case sensitivity to interferences is lower.

Another important advantage of this variant is that the energy of thepeaks 214 is much more spread for the higher order harmonics than forthe fundamental. Importantly, the sensitivity to the capacity variationsis essentially independent from the modulation frequency fmod, becausethe signal Vout is detected synchronously. Therefore, the proposedinvention provides a better rejection of interferences withoutcompromising the sensitivity tiny capacity changes.

In the above-mentioned example, only five different modulationfrequencies were used. It should be understood, however that themodulation frequency could be varied in whichever manner, either byhopping between a suitable number of discrete frequencies, or in acontinuous fashion.

Preferably, the modulation frequencies are optimized: very lowfrequencies should be avoided because they would significantly increasethe averaging time and hence reduce the frame rate. Very high modulationfrequencies should also be avoided because the performance would then belimited by parasitic time constants, by the speed of the chargeamplifier and by that of the ADC. Moreover operating at a too highmodulation frequency could require a higher current consumption.

In a preferred embodiment the modulation frequency is selected tooptimise averaging time and parasitic time constants i.e. the frequencyof the modulation voltage is varied only with modulation frequencieswhich are not so low as to cause a significant increase the averagingtime, and not so high so that performance would then be limited byparasitic time constants. In this embodiment the modulation frequency isvariable within a given range between a predefined minimum frequency(fmod_min) and a predefined maximum frequency (fmod_max).

In another more preferable embodiment, the difference between twoconsecutive modulation frequencies is optimized so that the differenceis greater than the width of the peaks of the transfer functioncorresponding to each frequency. In the example illustrated in FIG. 11,it can be seen that the difference between consecutive modulationfrequencies is greater than the width of the peaks of the transferfunction corresponding to each frequency. The difference “D” between themodulation frequencies fmod3 and fmod2, for example, is greater than thewidth “W” of the peaks 213, 212 in the respective transfer functionscorresponding to modulation frequency fmod3 and fmod2. This ensures thatthe peaks of the transfer functions corresponding to each of thedifferent frequencies do not overlap. Advantageously this leads to astrong improvement in the overall transfer function when averaging orperforming synchronous detection. It should be understood that the widthof a peak is measured at the widest portion of the peak.

Thus, in a preferable embodiment of the present invention, the spreadingof the peak in the transfer function will be significantly improved aslong as the number of averaged modulation frequencies remains below(fmod_max−fmod_min)/W, for the peak around fundamental, andk×(fmod_max−fmod_min)/W for the k-th order harmonics. Selecting a highernumber of modulation frequencies, the additional advantage will tend tosaturate, because the individual transfer functions become morecorrelated, having the peaks more overlapping.

However, there is also no significant drawback in varying the modulationfrequency on a given frequency range with a finer step, by selectingmore frequency points. The modulation frequency can namely be selecteddifferent for each modulation cycle, or even between two consecutivesampling of input capacitor, or else swept continuously during theoverall averaging time, or measurement time.

There is therefore an infinite number of possible ways of varying themodulation frequency (fmod) of the modulating voltage during theaveraging time or between successive synchronous detections. Forexample, but not limited exclusively to these examples, in the presentinvention:

-   -   The modulation frequency can vary by step or continuously, or        quasi continuously (with very small steps).    -   The modulation frequency can be varied randomly or pseudo        randomly, or according to a predetermined algorithm.    -   The variation of the modulation frequency can be identical or        different for different measurement cycles.    -   The variation of the modulation frequency can be a periodic        function. In such a case, the period of the variation of the        modulation frequency can be the same as the measurement        frequency (frame rate) or different. In the first case, the        modulation frequency will vary the same way for all the        measurements, while in the second case not. Preferable, the        periodicity of the modulation frequency at least more or less        corresponds to the measurement period.

Different periodic functions that can be used to vary the modulationfrequency (fmod) of the modulating voltage are shown in FIGS. 12a-e . Inparticular, FIGS. 12a-e shows different periodic functions that can beused to vary the instantaneous modulation frequency (fmod_inst) of themodulating voltage between predefined limits fmod_min and fmod_maxduring the averaging time or between successive synchronous detections.

In the graph illustrated in FIG. 12a , the instantaneous frequencyfmod_inst follows a saw-tooth pattern, while FIG. 12b shows another formof linear shift, fmod_inst being swept according to a triangle function.In the graph illustrated in FIG. 12c fmod_inst is swept according to acosinusoidal function. In the graph illustrated in FIG. 12d , fmod_instis varied step-wise. FIG. 12d shows an example of a periodic modulationwhose period does not coincide with the averaging time.

It should be understood that FIGS. 12a-e provide non-limiting examplesof the possible manner in which the modulation frequency (fmod) of themodulating voltage can be varied during the averaging time or betweensuccessive synchronous detections. It must also be understood thatmodulation frequency (fmod) of the modulating voltage can be varied inany other suitable manner; an infinite number of other functions arepossible.

In any embodiment of the present invention, the variation of themodulation frequency (fmod) of the modulating voltage can be achieved byany suitable means. As a non-limiting example, the variation of themodulation frequency (fmod) of the modulating voltage can be achieved bychanging the division ratio of a clock divider in order to adapt themodulation period; and/or by acting on the frequency of an oscillator,for instance trimming of RC oscillator, control of a Voltage ControlledOscillator (VCO).

Various modifications and variations to the described embodiments of theinvention will be apparent to those skilled in the art without departingfrom the scope of the invention as defined in the appended claims.Although the invention has been described in connection with specificpreferred embodiments, it should be understood that the invention asclaimed should not be unduly limited to such specific embodiment.

The invention claimed is:
 1. A measuring circuit electrically connectedto a capacitive touch-sensitive panel, the capacitive touch-sensitivepanel including a plurality of sense electrodes, the measuring circuitbeing adapted to electrically connect to the sense electrodes, andmeasure variations in an instantaneous electric capacity of the senseelectrodes in response to proximity to conductive bodies, wherein thesense electrodes are biased at a fixed voltage relative to a commonfloating reference potential, the measuring circuit comprising: a powermanagement circuit comprising a modulated voltage source generating amodulation voltage having a variable frequency that determines saidcommon floating reference potential, one or more readout units, eachconnected to a subset of the sense electrodes and operatively arrangedfor generating signals representing the instantaneous electric capacityof the sense electrodes in the subset; a detector for generating saidsignals synchronously with the modulation voltage.
 2. The measuringcircuit of claim 1, wherein the common floating reference potential isavailable at a guard terminal of the measuring circuit that iselectrically connected to a common guard electrode of the capacitivetouch-sensitive panel.
 3. The measuring circuit of claim 1, the detectorincluding an analog/digital converter configured to sample an inputsignal at a frequency which is synchronized to the variable frequency ofthe modulating signal; and a filter arranged to average those samplesover an averaging interval, wherein the averaging interval is a timeinterval over which a predefined number of samples have to be taken. 4.The measuring circuit of claim 3 wherein variable frequency of themodulation voltage changes ‘N’ times within the averaging interval,wherein ‘N’ is an integer value greater than ‘2’ so that the modulationvoltage has at least two different modulation frequencies within theaveraging interval.
 5. The measuring circuit of claim 4 wherein the ‘N’frequencies to which frequency of the modulation voltage is varied, areall within a predefined frequency range.
 6. The measuring circuit ofclaim 4 wherein the difference between two frequencies of the modulationvoltage before and after a change is such that peaks of the transferfunctions corresponding to each frequency do not overlap.
 7. Themeasuring circuit of claim 3 wherein the means for varying the frequencyof the modulation voltage is configured to vary the frequency of themodulation voltage in at least one of the following manners within theaveraging interval: continuously; randomly; pseudo randomly; accordingto a predefined algorithm; and/or according to a periodic function.
 8. Amethod for measuring variations in an instantaneous electric capacity ofsense electrodes in a touch-sensitive panel, in response to proximity toconductive bodies, wherein the sense electrodes are biased at a floatingreference voltage, the touch-sensitive panel including a common guardelectrode, the method comprising the steps of generating a modulationvoltage that determines said floating reference voltage with a voltagesource comprised in a power management circuit, detecting theinstantaneous electric capacity of the sense electrodes synchronouslywith the modulation voltage, generating signals representing theinstantaneous electric capacity of the sense electrodes; varying afrequency of the modulation voltage.
 9. The method of claim 8 whereinthe step of including sampling an input signal at a frequency which issynchronized to the frequency of the modulation voltage; and averagingthose samples over an averaging interval, wherein the averaging intervalis a time interval over which a predefined number of samples have to betaken.
 10. The method of claim 9 including varying the frequency of themodulation voltage ‘N’ times within the averaging interval, wherein ‘N’is an integer value greater than ‘2’ so that the modulation voltage hasat least two different modulation frequencies between successivesynchronous detections.
 11. The method of claim 10 wherein the frequencyof the modulation voltage is varied within a predefined frequency range.12. The method of claim 10, including varying the frequency of themodulation voltage to provide a difference between two differentfrequencies of the modulation voltage before and after a change whichensures that peaks of a transfer functions corresponding to eachfrequency do not overlap.
 13. The method of claim 12 including varyingthe frequency of the modulation voltage so that the difference betweentwo different frequencies of the modulation voltage before and after achange is greater than a broadest width of peaks of the transferfunctions corresponding to each of the two different frequency, suchthat peaks of the transfer functions corresponding to each frequency donot overlap.
 14. The method of claim 9, including varying the frequencyof the modulation voltage in at least one of the following mannerswithin the averaging interval: continuously; randomly; pseudo randomly;according to a predefined algorithm; and/or according to a periodicfunction.
 15. The method of claim 8 comprising changing the divisionratio of a clock divider and/or adjusting the frequency of anoscillator.